JPH08191224A - Mobile communication equipment - Google Patents

Abstract

PURPOSE: To reduce the size, weight and power consumption of this mobile communication equipment by allowing drain voltage to be supplied to a power amplifier constituting a transmission part in accordance with a transmission power control signal. CONSTITUTION: The transmission part 32 is provided with a gate voltage setting circuit 323 and a drain voltage setting circuit 324 and the gate voltage VG and drain voltage VD of the power amplifier 322 are variably set up by a power control signal PCS received from a control circuit 20 through a time- division multiplexing/separating circuit 23. In the case of reducing transmission power from the maximum power in accordance with transmission power control, the input power of the amplifier 322 is reduced by a variable attenuator 321 in accordance with the power control signal PCS outputted from the circuit 20. In this case, the drain voltage VD of the amplifier 322 is also dropped to within a fixed voltage range by a drain voltage setting circuit 324 in accordance with the signal PCS. Consequently the efficiency of the power amplifier 322 at the time of maximum transmission can be held also at the time of controlling transmission power.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a mobile communication device, and more particularly to a device such as a mobile communication terminal used in a mobile communication system such as a car phone or a mobile phone.

[0002] In recent years, mobile phones, which can be contacted with a desired party at any time regardless of location, have been remarkably spread. However, in the future, the demand for easy mutual communication at home, office, factory, etc. will increase the demand. Is expected to increase further. In mobile communication devices such as mobile phones,
It is necessary to reduce the power consumption, reduce the size and weight, and reduce the price.

[0003]

2. Description of the Related Art FIG. 14 is a diagram showing the overall configuration of a conventionally proposed digital car telephone terminal (digital mobile device). When transmitting / receiving to / from the base station, this digital mobile device can switch the magnitude of the transmission power in six steps based on an instruction from the base station so that the received electric field strength at the base station can be kept within a certain range. It is configured to be able to. The transmission system uses a so-called direct modulation system that directly converts a baseband signal near 21 kHz into a transmission RF signal near 950 MHz, while the reception system uses a double superheterodyne reception system. Hereinafter, each circuit constituting this digital mobile device will be described.

In FIG. 14, a control circuit 20 is composed of a microcomputer and is a circuit for communicating with the control circuit of the telephone body and for controlling incoming and outgoing call connection on the terminal side including channel switching control and location registration control. . The tone generation circuit 21 is a circuit that generates a tone signal based on the transmission control data from the control circuit 20. The encoding / decoding circuit 22 is a circuit that encodes a transmission analog audio signal input from a microphone into encoded audio data, or converts the received encoded audio data into an analog signal and outputs the analog signal to a speaker.

Time division multiplexing / demultiplexing circuit (TDMA circuit) 2
3 time-division-multiplexes encoded voice data and transmission control data from the control circuit 20 by the TDMA method, and outputs transmission baseband data at a desired transmission time slot designated by the control circuit 20, It is a circuit for separating encoded voice data and reception control data relating to a desired reception time slot from reception baseband data.

The transmission signal processing circuit 30 performs serial / parallel conversion, π / 4 shift, etc., on the transmission baseband data and outputs it.
This is a circuit for outputting 1 kHz I-channel and Q-channel digital signals. The quadrature modulator 31 uses the first local oscillation signal (f _L1 ) from the PLL synthesizer circuit 50 on the I channel and Q channel data processed by the transmission signal processing circuit 30 to perform π / 4 shift QPSK burst modulation. Is a circuit for converting the signal into a 950 MHz π / 4 shift QPSK radio frequency (RF) signal and outputting the converted signal.

The PLL synthesizer circuit 50 is a circuit for outputting a first local oscillation signal for transmission and reception (f _L1 : near 950 MHz) whose frequency changes according to frequency variable control accompanying the channel switching control of the control circuit 20. .
FIG. 17 shows a detailed configuration example of the PLL synthesizer circuit 50. As shown in the figure, the first PLL synthesizer circuits 511 and 7 which output oscillation signals in the vicinity of 200 MHz respectively according to the frequency variable control from the control circuit 20.
A second PLL synthesizer circuit 512, which outputs an oscillation signal near 50 MHz, and first and second PLL synthesizer circuits 5
A frequency converter 513 for generating a first local oscillation signal in the vicinity of 950 MHz based on the output signals of 11, 512, a hybrid circuit 514 for branching the output signal of the frequency converter 513 into a transmission side and a reception side, and a transmission system. An amplifier 515 for adjusting the magnitude of the first local oscillation signal, an amplifier 516 for adjusting the magnitude of the first local oscillation signal to the receiving system, and the like are included.

The transmission unit 32 transmits the transmission R from the quadrature modulator 31.
It is a circuit that adjusts the power of the F signal according to an instruction from the control circuit 20, and includes a variable attenuator 321, a power amplifier 322, and the like. The variable attenuator 321 controls the power control signal P designated by the control circuit 11 via the TDMA circuit 23.
Power amplifier 3 by CS (3-bit digital signal)
This is a circuit that sets 22 inputs in 6 steps with a 4 dB step width. The power amplifier 322 is a high-efficiency power amplifier using a GaAs FET, and the gate voltage VG and the drain voltage VD are set to fixed values.

FIG. 16 shows a circuit configuration example of the above variable attenuator. As shown in the figure, it does not include active elements such as transistors, but consists of resistors, capacitors, diodes, etc., and the attenuation rate is accurately 4 dB steps according to the power control signal (analog signal) input to the control power input terminal CONT. The width can be changed in 6 steps. In FIG. 14, the transmission system circuit is a D / A that D / A converts the power control signal PCS (digital signal) from the time division multiplexing / demultiplexing circuit 23 into an analog signal on the control signal input side to the variable attenuator.
Although a converter is included, it is omitted in FIG. 14.

FIG. 15 shows a circuit configuration example of the above-mentioned power amplifier 322. This power amplifier is a two-stage amplifier circuit composed of transistors TR1 and TR2, as shown in the figure,
The drain voltage VD applied to the drain side of the transistors TR1 and TR2 and the gate voltage VG applied to the gate side of the first-stage transistor TR1 are the power amplifier 32.
It is fixedly set to a fixed value that maximizes the efficiency of the amplifier with respect to the maximum amplitude value of the input signal P _in . The rectangular block in the figure represents an inductor formed by a circuit pattern.

The transmission / reception branching filter 33 is a circuit for guiding the power-transmitted transmission RF signal output from the transmission section 32 to the transmission / reception antenna 35 and the reception RF signal received by the antenna 35 to the reception system. Yes, it includes a transmission band filter, a reception band filter, and a combiner.

The low noise RF amplifier 49 is a circuit for amplifying the received RF signal received by the antenna 35. The bandpass filter 48 is a circuit that passes only a predetermined RF band of the received RF signal amplified by the low noise RF amplifier 49. The first frequency converter 47 is a circuit that converts the received RF signal that has passed through the band-pass filter 48 into a first intermediate frequency signal of a predetermined frequency using the receiving first local oscillation signal from the PLL synthesizer circuit 50. The first intermediate frequency filter 46 is a circuit that extracts the first intermediate frequency signal from the output signal of the first frequency converter 47.

The local oscillator 51 is a second oscillator having a predetermined fixed frequency.
This is a circuit that outputs a local oscillation signal (f _L2 ). The second frequency converter 45 is the first frequency converter 45 after passing through the first intermediate frequency filter 46.
It is a circuit that converts the intermediate frequency signal into a second intermediate frequency signal of a predetermined frequency using the second local oscillation signal from the local oscillator 51. The second intermediate frequency filter 44 is a circuit that extracts the second intermediate frequency signal from the output signal of the second frequency converter 45. The intermediate frequency amplifier 43 includes a second intermediate frequency filter 44.
It is a circuit that amplifies the second intermediate frequency signal after passing. The limiter amplifier 42 is a circuit that limits the amplitude of the second intermediate frequency signal.

The quadrature demodulator 41 is a circuit for performing π / 4 shift QPSK burst demodulation on the burst output signal from the limiter amplifier 42. The reception signal processing circuit 40 is a circuit that performs parallel-serial conversion or the like on the I-channel and Q-channel demodulated signals demodulated by the quadrature demodulator 41 to generate a reception baseband signal and outputs the reception baseband signal to the time division multiplexing / demultiplexing circuit 23. . The electric field strength detection circuit 52 is a circuit that detects the received electric field strength based on the second intermediate frequency signal extracted from the intermediate frequency amplifier 43 and outputs a detection signal to the control circuit 20.

FIG. 18 shows another example of the configuration of a conventional mobile unit for the digital mobile communication system. This digital mobile device is of the TDMA system, and uses a so-called indirect modulation system in which a transmission system once converts a baseband signal into an intermediate frequency signal and then up-converts it to a transmission RF signal. The receiving system uses the double superheterodyne receiving system.

In FIG. 18, a control circuit 51 is a circuit that controls the entire apparatus including a microprocessor. The modulation baseband circuit 52 corresponds to the above-mentioned transmission signal processing circuit and generates a transmission baseband signal. The transmission baseband signal from the modulation baseband circuit 52 is output from the local oscillator 71 by the modulator 53.
A local oscillation signal near MHZ is used for digital modulation into a modulation signal near 82 MHZ. This modulated signal is further transmitted by the frequency converter 54 from the PLL circuit side at 1365 MHz.
Transmission R near 1446MHz using local oscillation signal
The F signal is up-converted, and the transmission RF signal is transmitted through the power amplifier 55, the transmission / reception duplexer 33, and the antenna 35.

On the other hand, the reception RF received by the antenna 35
The signal passes through the duplexer 33, is amplified by the low noise amplifier 61, and is then input to the frequency converter 62 to be converted into a first intermediate frequency signal using the first local oscillation signal in the vicinity of 1365 MHz from the distributor. Then, it is input to the frequency converter 63, where it is converted into a second intermediate frequency signal using the second local oscillation signal in the vicinity of 129.5 MHz from the local oscillator 70, and is digitally demodulated by the demodulator 65.

The circuit that supplies the local oscillation signal to the frequency converter 54 on the transmitting side and the frequency converter 62 on the receiving side is a PLL.
It is composed of a circuit including a synthesizer circuit. That is, a first PLL synthesizer circuit 66 that outputs an oscillation signal near 200 MHz corresponding to the channel setting data, a second PLL synthesizer circuit 67 that outputs an oscillation signal near 1165 MHz corresponding to the channel setting data, both synthesizer circuits 66 and 67. It includes a frequency converter 68 that outputs a local oscillation signal in the vicinity of 1365 MHz based on the output signal of 1), and a distributor 69 that branches the local oscillation signal output from the frequency converter 68 into a transmission side and a reception side.

FIG. 19 shows an example of the configuration of this PLL synthesizer circuit. This configuration example is general as a PLL synthesizer circuit. In FIG. 19, the reference crystal oscillator 66 generates a reference signal having a reference frequency, and the reference signal is frequency-divided by the prescaler 6611 of the PLL unit 661 and the phase comparator 6613.
On the other hand, the output oscillation signal which has been fed back and is fed back is divided by the prescaler 665 and then inputted to the other input terminal of the phase comparator 6613 through the 1 / n programmable counter 6612. The phase comparator 6613 compares the phases of both input signals and inputs the output signal of the comparison result to the voltage controlled oscillator (VCO) unit 664 as a control voltage through the charge pump 662 and the loop filter 663. The voltage controlled oscillator circuit 664 outputs an output signal having an oscillation frequency corresponding to the input control voltage, and part of it is fed back to the prescaler 665 side described above. The voltage controlled oscillator circuit 664 controls the oscillation frequency of the output signal by changing the LC resonance frequency by changing the capacitance of the diode VD according to the control voltage applied to the variable capacitance diode VD. Therefore, according to the channel setting data from the control circuit 51, the 1 / n programmable counter 661
By changing the division ratio of 2, the voltage controlled oscillation circuit 66
4 can control the frequency of the oscillation signal.

[0020]

In the above-mentioned conventional mobile communication device, it is necessary to reduce the size and weight of the device, reduce power consumption, and reduce cost. This will be described below. For example, in the case of a mobile communication device such as a mobile phone, a battery is used as a power source to operate the mobile communication device because of its portability. However, as a matter of course, a call cannot be made if the battery capacity is exhausted. Therefore, when making a call using a battery having a limited capacity, reducing the current consumption of the power amplifier, which consumes a large amount of current in the device, is a key point for extending the call time of the mobile phone.

On the other hand, in the conventional communication terminal as shown in FIG. 14, the transmission power control is performed by the variable attenuator 321 provided in front of the power amplifier 322.
22 has a constant amplification factor, and its drain voltage VD and gate voltage VG have a certain fixed value regardless of the transmission power (that is, a value at which the amplifier has the maximum efficiency for the maximum amplitude value of the input signal). ) Is fixed.

As a result, as compared with the efficiency of the power amplifier at the time of maximum transmission, as the transmission power is controlled and the transmission power is lowered, the power amplifier operates away from the saturation point and the power of the amplifier is increased. Efficiency deteriorates. That is, the power amplifier consumes more power than necessary even when the amplitude of the input signal is small. Therefore, it can be said that there is a loss in the power efficiency of the entire device, but the life of the battery is shortened.

At present, as a mobile phone terminal, TDM is used.
Digital mobile phones that have adopted the digital modulation / demodulation method of A have been put into practical use, but this method tends to be expensive because it essentially has more functional block circuits than mobile phone terminals that use the analog modulation / demodulation method. Therefore, it is a future task to reduce the price of analog machines. Therefore, each functional block circuit used in the device is required to have a circuit configuration that allows the use of a low cost.

Considering the device circuit configuration from this viewpoint, there are the following problems. That is, the π / 4 shift QPSK modulation method applied to the digital portable device is
Since it is a linear modulation system, the transmission circuit is required to have linearity without distortion. For this reason, as an RF gain variable element for controlling the transmission power of the transmission section, use of a variable amplifier often has difficulty in distortion characteristics and the like, and usually the distortion characteristics do not change even if gain control is performed. Thus, the variable attenuator 321 is used. Further, as the variable attenuator 321, a variable attenuator capable of accurately switching the attenuation rate in 5 steps with a 4 dB step width is currently required, but the variable attenuator 321 is expensive. Further, since the variable attenuator 321 is controlled in the attenuation direction and has no amplification function, it is necessary that a signal of a required magnitude corresponding to the maximum transmission power is input at the input end position of the variable attenuator 321. Therefore, there is a problem that a driver amplifier for amplifying the output signal of the quadrature modulator 31 to a required size is further required before the variable attenuator, resulting in an increase in the cost of the device.

Further, in the conventional mobile device, the PLL synthesizer circuit 50 is used as a circuit for supplying the local oscillation signal (f _L1 ) to the transmitting system circuit and the receiving system circuit. As shown in 17,
Since one local signal output from the frequency converter 513 is distributed by the hybrid circuit 514 for transmission and reception, the level drops. Therefore, the amplifiers 515 and 516 provide voltage levels suitable for the transmission circuit and the reception circuit, respectively. Is amplified up to. Therefore, there is a problem in that the power consumption is increased, the circuit scale is increased, and the manufacturing cost is increased by the amount corresponding to the amplifiers 515 and 516.

In the case of an indirect modulation type communication device as shown in FIG. 18, local oscillators 71 and 70 for modulation and frequency conversion are required on the transmitting side and the receiving side, respectively. Since the transmitting side and the receiving side have the local oscillators independently as described above, the number of circuit components is increased, which is a factor that hinders low power consumption, small size and light weight, and low price.

The present invention has been made in view of the above problems, and it is an object of the present invention to reduce the size and weight of a mobile communication device, reduce the power consumption, and reduce the price.

[0028]

FIG. 1 is an explanatory view of the principle according to the present invention. In order to solve the above-mentioned problems, the mobile communication device according to the present invention is, as one form,
A transmission unit 84 that switches and sets the transmission power in multiple stages according to the transmission power control signal, and a drain voltage supplied to the power amplifier 83 that configures the transmission unit 84 is switched and set in stages according to the transmission power control signal. And a drain voltage setting circuit 85 for controlling the drain voltage.

According to such a configuration, when the magnitude of the transmission power is switched, the drain voltage of the power amplifier 83 of the transmission section 84 is also switched to a value having a power efficiency suitable for the transmission power, so that the power consumption is reduced. Efficiency is improved as a whole, and low power consumption can be achieved.

The above-mentioned mobile communication device may be configured such that the drain voltage is switched and set by the drain voltage setting circuit in several steps on the side of the higher transmission power in the transmission power control. Thus, it is possible to reduce the circuit scale of the drain voltage setting circuit, and to reduce the size, weight, and cost.

Further, the above-mentioned mobile communication device may further include a gate voltage setting circuit for switching and setting the gate voltage to be supplied to the power amplifier forming the transmitting section in multiple stages according to the transmission power control signal. Of course, with this configuration, the power efficiency of the power amplifier can be made more appropriate according to the transmission power at that time.

Further, in the mobile communication device described above, when the power amplifier has a multi-stage amplifier configuration, the drain voltage setting circuit may be provided only in the amplification stage having a large inflow current. The circuit scale of the drain voltage setting circuit can be reduced and the characteristic condition of the power amplifier can be relaxed.

Further, in the mobile communication device described above, when the power amplifier has a multi-stage amplifier configuration, each amplification stage may be provided with a drain voltage setting circuit. By controlling, it is possible to further improve the power efficiency.

As another mode, the mobile communication device according to the present invention may have a modulator 81 for modulating a transmission baseband signal into an intermediate frequency band by a predetermined modulation method, and a modulated wave output of the modulator 81 as desired. A frequency converter 82 for converting to a radio transmission wave frequency, a local oscillation circuit for generating a local oscillation signal used for frequency conversion in the frequency converter 82, and a gain of the local oscillation signal of the local generation circuit for a transmission power control signal. Accordingly, the gain variable element 87 is provided that is switched stepwise and supplied to the frequency converter.

With this configuration, during transmission power control, the frequency converter 8 is passed through the variable gain element 87.
The amplitude of the local oscillation signal supplied to 2 can be changed according to the transmission power control signal, whereby the power of the transmission signal supplied to the transmission unit 84 can be changed, and thus the transmission power can be changed in multiple stages. it can.

This mobile communication device is provided with a second gain variable element before or after the frequency converter, and the gain of the gain variable element is switched stepwise according to the transmission power control signal to change the gain. The transmission power may be switched in multiple stages by the combination of the element and the second gain variable element, and by such a configuration, the gain variable ability of the gain variable element is shared with the second gain variable element. Therefore, the characteristics of the device can be relaxed.

Also, this mobile communication device is configured to switch the amplitude value of the transmission baseband stepwise according to the transmission power control signal, and to switch the transmission power in multiple stages by combining this with the gain variable element. Alternatively, with such a configuration, the variable gain element 87 and the variable gain capability can be shared, and the characteristics of the variable gain element can be relaxed.

Further, the mobile communication device further includes a drain voltage setting circuit for setting the drain voltage supplied to the power amplifier for amplifying the output signal of the frequency converter in a stepwise manner according to the transmission power control signal. The gain variable element 86 can be shared with the gain variable element 86, and the characteristics of the gain variable element can be relaxed.

Further, this mobile communication device may be configured to control the gain variable element 86 to the maximum attenuation state or the operation stop at the timing when transmission is not performed. It is possible to obtain the effect of suppressing the emission of unnecessary waves at the off timing of.

As another mode, the mobile communication device according to the present invention comprises a modulator for modulating a transmission baseband signal by a predetermined modulation method, and a local oscillation signal used for the modulation operation in the modulator. A local oscillator circuit that is generated and supplied to the modulator; and a switcher that switches the output signal of the modulator to the transmission power amplifier side during transmission and to the frequency converter side of the reception circuit during reception, At the time of reception, the supply of the transmission baseband signal to the modulator is stopped to make the modulator output a non-modulated wave.

With this configuration, at the time of transmission, the transmission baseband signal supplied to the modulator is modulated by the output signal of the local oscillation circuit and transmitted as a modulated wave.
During reception, the output signal of the modulator can be used as a non-modulated wave and used as a local oscillation signal for frequency conversion on the receiving circuit side. Therefore, a branch circuit and an amplifier for amplifying the output thereof are not required, so that the circuit scale and power consumption can be reduced and the cost can be reduced.

In another form, the mobile communication device according to the present invention is a mobile communication device which performs communication by shifting the transmission timing and the reception timing, and includes first and second PLL synthesizer circuits. Of the first PLL synthesizer circuit, the local oscillation circuit including a frequency conversion circuit for generating a combined local oscillation signal for performing frequency conversion from the output signal of By using the oscillation output signal as the local oscillation signal for modulation of the transmitter modulator and changing the frequency of the oscillation output signal of either the first or second PLL synthesizer circuit during transmission and reception, The frequency of the combined local oscillation signal is changed so as to match the frequency conversion of the transmission wave and the intermediate frequency conversion of the radio reception wave.

With this configuration, an independent local oscillation circuit is not required to supply the local oscillation signal for modulation to the transmission side modulator, so that the circuit scale and power consumption can be reduced and the cost can be reduced. Can be planned.

In the mobile communication device described above, a capacitor is added in parallel with a variable capacitance diode for changing the oscillation frequency of the voltage controlled oscillation circuit to the PLL synthesizer circuit for changing the frequency at the time of transmission and at the time of reception. The varactor diode may be connected or disconnected at the time of transmission and at the time of reception by the switch means. With this configuration, it is possible to instantly switch the capacitance of the variable capacitance diode portion at the time of switching the oscillation frequency and quickly stabilize the oscillation frequency to a desired oscillation frequency.

In the mobile communication device, the PLL synthesizer circuit that changes the frequency during the transmission and the reception has an application circuit that applies a bias voltage to the variable capacitance diode that changes the oscillation frequency of the voltage controlled oscillation circuit. It may be configured to change the Baice voltage of the applying circuit at the time of transmission and at the time of reception, and by such a configuration, the capacitance of the variable capacitance diode portion is instantaneously switched at the time of switching the oscillation frequency, and It can be stabilized at a desired oscillation frequency.

[0046]

Embodiments of the present invention will be described below with reference to the drawings. Throughout the following drawings, circuit blocks having the same functions are designated by the same reference numerals.

FIG. 2 shows the overall configuration of a mobile communication terminal as an embodiment of the present invention. In this embodiment, the power efficiency of the power amplifier 322 is improved to reduce the power consumption. In the drawing, the same circuit blocks as those described in the section of the related art are designated by the same reference numerals. The difference is that the transmission unit 32 includes a gate voltage setting circuit 323 and a drain voltage setting circuit 324, and the power amplifier 3
This is that the gate voltage VG and the drain voltage VD of 22 can be variably set by the power control signal PCS received from the control circuit 20 via the time division multiplexing / demultiplexing circuit 23.

FIG. 3 shows a configuration example of the drain voltage setting circuit 324. The gate voltage setting circuit 323 is set to D
It is configured using a C / DC converter circuit. The drain voltage setting circuit 324 is configured so that the drain voltage VD supplied to the power amplifier 322 can be switched to a plurality of fixed values VD1, VD2, VD3 ... In response to the transmission power control signal PCS.

In FIG. 3, the transmission power control signal PCS
The (3-bit digital signal) is input to the power control signal decoder 3241 and decoded into a signal indicating which of the plurality of drain voltage values to switch to,
This decoded signal is input to the reference voltage source 3242. The reference voltage source 3242 generates a reference fixed voltage corresponding to the drain voltage indicated by the input signal, and the error amplifier 324.
3 (+) terminal. The error amplifier 3243 drives the base of the transistor 3244 so that the drain voltage output on the collector side of the transistor 3244 is fed back to the (−) terminal by the feedback circuit 3245 and is controlled so that the drain voltage output becomes constant. With this configuration, the drain voltage setting circuit 324 generates one of the plurality of drain voltages VD1, VD2, VD3 ... According to the value of the power control signal PCS, and supplies it to the power amplifier 322. be able to.

The drain voltage setting circuit 324 is not limited to that of this embodiment, and may be, for example, one based on a regulator or one based on a DC / DC converter. .

The operation of the apparatus of this embodiment will be described below.
When the transmission power is reduced from the maximum power according to the transmission power control, the input power of the power amplifier 322 is reduced by the variable attenuator 321 according to the power control signal from the control circuit 20. In this case, the drain voltage setting circuit 324 also drops the drain voltage VD of the power amplifier 322 in a certain voltage range according to the power control signal PCS. This value is determined by the single characteristic of the power amplifier 322. The purpose of decreasing the drain voltage in this way is to maintain the power efficiency of the power amplifier 322 at the time of maximum transmission even when the transmission power control is performed.

FIG. 4 shows the power for each drain voltage value.
Input / output characteristics of the amplifier 322 are shown. In and out of this Figure 2
To maintain the power efficiency of the above power amplifier using force characteristics
explain about. In Fig. 4, the characteristics shown on the upper side are
Input power P_inOutput power at each drain voltage VD
_outIs the characteristic of the input power P _in
The characteristics of the current consumption ID at each drain voltage VD for
is there. As shown in FIG. 4, for example, the maximum input power P_inTo 1
On the other hand, when VD1 is selected as the drain voltage VD,
Output power is P_out1 and power consumption at that time
The flow is ID1. Input power P here_inIs P_inReduced to 3
VD1 is used as it is as the drain voltage VD
Output power P_outHas good linearity and causes distortion
Although it does not exist, ID1 "is required as current consumption.
Input power P_inIs P_inDrain in accordance with the decrease to 3
The drain voltage is reduced to VD3 by the voltage setting circuit 324.
If so, the output power at that time is P_outThis operating point in 3
The linearity at is kept good, while the current consumption is ID3.
Since ID1 ″> ID3, the power amplifier 322
Can reduce current consumption and improve power efficiency
It

When the transmission power control is performed in this way, by changing the drain voltage accordingly, deterioration of the power efficiency of the power amplifier 322 can be suppressed, and the transmission power control can be performed as in the actual use condition of the portable device. Is frequently performed in the system, the power efficiency of the transmitting section as a whole is improved and the power consumption is reduced. As a result, the battery life is extended and the talk time is extended.

In this embodiment, since the drain voltage to be set is a constant value for each transmission power, the efficiency improvement effect at maximum transmission is inferior to the method of dynamically controlling the drain voltage according to the transmission information. However, since it has little influence on the distortion characteristic of the power amplifier 322, there is an advantage that the drain voltage setting unit can be a simple and inexpensive circuit.

It should be noted that the number of variable drain voltage switching stages in the drain voltage setting circuit 324 does not necessarily have to match the number of variable attenuator 321 switching stages, and the drain voltage can be varied only in a few steps on the high transmission power side. The drain voltage setting circuit 324 may be configured to reduce the circuit scale of the drain voltage setting circuit 324.

In this embodiment, the drain voltage of both the two-stage transistors TR1 and TR2 of the power amplifier shown in FIG. 15 is variable, but in the case where the power amplifier has a multi-stage configuration as described above. In this case, it is not always necessary to make the drain voltages of the transistors of all the stages variable, and for example, the drain voltage of only the stage having a large inflow power (for example, the transistor TR2 on the rear stage side) can be variably set. The circuit scale may be reduced and the characteristic conditions of the power amplifier may be relaxed. Further, even when the drain voltages of the transistors in all the stages are varied, a drain voltage setting circuit may be independently provided for each stage to perform more accurate control and further improve the efficiency.

Similarly to the above, the gate voltage setting circuit 323 also sets the gate voltage VG of the power amplifier 322 to a predetermined plurality in accordance with the power control signal (that is, according to the magnitude of the input power P _{in to} the power amplifier 322). By variably setting any of the fixed gate voltage values, the current consumed by the power amplifier 322 is set to a large value when the transmission power is high and a small value when the transmission power is small. The efficiency can be kept within a certain range regardless of the input power.

FIG. 5 shows another embodiment of the present invention.
This embodiment is intended to reduce the cost of the device by using a circuit configuration that does not require expensive elements such as a variable attenuator, which is required in the high frequency part of the conventional circuit. In the figure, as a difference from the above-described embodiment apparatus, the embodiment apparatus of FIG. 2 uses a direct modulation method as a modulation method, whereas this embodiment circuit uses an indirect modulation method. Therefore, the 21 kHz baseband signal from the transmission signal processing circuit 30 is used in the intermediate frequency quadrature modulator 311 by using the 200 MHz local oscillation signal from the PLL synthesizer circuit 50 ′ to generate a 200 MHz π signal.
After being modulated into an intermediate frequency signal of / 4 shift QPSK, this intermediate frequency signal is input to the up-conversion mixer 312, and this up-conversion mixer 312 uses the local oscillation signal of 750 MHz from the variable gain element 34 to generate 95
The frequency is converted into a transmission RF signal of 0 MHz, and this is input to the power amplifier 322 without passing through the variable attenuator.

This intermediate frequency quadrature modulator 311 receives a constant carrier wave (200M) from the PLL synthesizer circuit 50 '.
Hz) is supplied, and the modulated baseband waves (I, Q signals) from the transmission signal processing circuit 30 are also adjusted to have a maximum output where the distortion of the intermediate frequency modulated wave spectrum is not deteriorated. It is like this.

FIG. 6 shows a circuit configuration example of the up-conversion mixer 312. This up-conversion mixer 3
Reference numeral 12 is a Gilbert cell type, and since these are balance types, a terminal IF and a power amplifier 32 to which the intermediate frequency signal from the intermediate frequency quadrature modulator 311 is input.
Terminal RF from which the RF signal after frequency conversion to 2 is output,
It has a terminal LO to which the local oscillation signal from the variable gain element 34 is input and respective inversion terminals. Further, the power supply voltage application terminal VS and the ground terminal GND are provided.

As the PLL synthesizer circuit 50 ',
In the circuit in FIG. 17, there is no circuit for supplying a local oscillation signal of 950 MHz to the transmission system circuit, but instead, the local oscillation signal (200 MHz) of the first PLL synthesizer circuit 511 is supplied to the intermediate frequency quadrature modulator 311, while the second PLL is supplied. The local oscillation signal (750 MHz) of the synthesizer circuit 512 is supplied to the variable gain element 34. The gain variable element 34 is a circuit that changes the amplitude value of the input local oscillation signal according to the control signal from the control signal generation circuit 36. The control signal generation circuit 36 has the same circuit configuration as the drain voltage setting circuit 324, and has the power control signal P
A control voltage having a different value is generated according to the value of CS.

FIG. 7 shows a circuit configuration example of the gain variable element 34. As shown, the transistor TR2 shunting a portion of the transistor TR1 and the input signal P _in for amplifying the input signal P _in the ground GND side
By changing the magnitude of the input signal current shunting to the transistor TR2 side in accordance with the magnitude of the control voltage input to the terminal CONT, the input passing from the input terminal to the output terminal as a result. Change the gain of the signal P _in .

The operation of the apparatus of this embodiment will be described below.
When the transmission power is reduced from the maximum power according to the transmission power control, the control signal generation circuit 36 lowers the output control voltage according to the power control signal PSC from the control circuit 20. As a result, the variable gain element 34 becomes the PLL synthesizer circuit 5
The power of the local oscillation signal supplied from 0 ′ to the up-conversion mixer 312 is reduced. Therefore, the output signal amplitude of the up-conversion mixer 312 also decreases in accordance with the drop in the power of this local oscillation signal. Therefore, the conventional variable attenuator 321 is not necessary if the gain of the variable gain element 34 is changed so that the output of the up-conversion mixer 312 has an output amplitude of 6 steps with a 4 dB step width in accordance with the power control signal PSC. Becomes

Comparing the device of this embodiment with the conventional device,
Conventionally, the transmission power control is performed by the variable attenuator 321 (or a variable gain element may be used) on the RF signal subjected to the π / 4 shift QPSK modulation. Therefore, the variable attenuator has good linearity and causes phase distortion and the like. However, in the present embodiment, since the transmission power control is performed on the local oscillation signal of the PLL synthesizer circuit 50 'which is an unmodulated carrier signal, The conditions can be relaxed with respect to the linearity and distortion characteristics required for the variable gain element 34, and a relatively inexpensive element can be used for the configuration.

In the embodiment of FIG. 5, the variable control of the amplitude value of the output signal of the up-conversion mixer 312 is performed only by the variable gain element 34 that changes the gain of the local oscillation signal, but the present invention is not limited to this. It is not limited.
For example, an element such as a variable gain element or a variable attenuator is further inserted between the intermediate frequency modulator 311 and the up-conversion mixer 312, and the variable amplitude value is controlled by a combination of these elements and the variable gain element 34. Alternatively, both elements can share the gain varying capability, and the characteristics required for the used element can be relaxed. In this case, the amplitude balance between the intermediate frequency signal amplitude input to the up-conversion mixer 312 and the local oscillation signal amplitude can be improved. Further, this newly inserted element may be one that can be switched in two steps at an operating frequency of around 200 MHz. Therefore, even if the gain adjustment of the π / 4 shift QPSK signal is performed, the conventional variable attenuator 321 can be used.
It is available at a low price compared to.

Further, the above-mentioned element is used as an intermediate frequency modulator 311.
Instead of being inserted between the up-converting mixer 312 and the up-converting mixer 312, it may be inserted between the up-converting mixer 312 and the transmitting unit 32. Further, the amplitude of the baseband signal generated in the transmission signal processing circuit 30 can be variably set according to the transmission power control, so that the combination of this and the gain variable element 34 can perform the amplitude control, and the gain variable element 321. It may be one that alleviates the gain variable ability required for the above.

Further, this stepwise control of the transmission power can be carried out in combination with the above-mentioned drain voltage control of the power amplifier 322. That is, as can be seen from the example of FIG. 4, since the power amplifier 322 can change the output power by changing the drain voltage, the drain setting voltage control of the drain voltage setting circuit 324 and the gain variable element 34 are used. Digital amplifier 322
If the final transmission power is controlled in combination with the input power control to, the characteristics of the variable gain element 34 can be relaxed and the above-mentioned power efficiency can be improved.

Further, in the embodiment of FIG. 5, it is unnecessary at the transmission OFF timing by setting the variable gain element 34 to the maximum attenuation state or stopping the function (output cutoff) at the timing when transmission is not performed. It is also possible to control so that no waves are sent out and obtain the effect of suppressing unnecessary waves.

FIG. 8 shows another embodiment of the present invention. This embodiment is intended to reduce the size and weight of the apparatus, reduce the power consumption, and reduce the cost by simplifying the circuit configuration of the PLL synthesizer circuit that is the source of the local oscillation signal. Hereinafter, this embodiment will be described with reference to the differences from the embodiment of FIG.

In FIG. 8, the PLL synthesizer circuit 5
0 "is the first and second PLL synthesizer circuits 511 and 51.
2. The hybrid circuit 513 and the amplifiers 514 and 51, which are composed of the frequency converter 513, as described in the related art.
5 has been deleted. 950 from the frequency converter 513
The local oscillation signal of MHZ is input to the quadrature modulator 31 and is not directly supplied to the receiving side. The output signal of the quadrature modulator 31 is supplied to the input of the transmission unit 32 through one of the switching contacts of the switch 517, and also passes through the other switching contact of the local oscillation signal to the first frequency converter 47 on the receiving side. Supplied as. The switch 517 is controlled to switch the output signal to the transmitter 32 side when the communication device is in the transmission mode, and to the first frequency converter 47 side when the communication device is in the reception mode. Further, the transmission signal processing circuit 30 is controlled so as to input the baseband signal to the quadrature modulator 31 in the transmission mode and to input a DC voltage in order to generate an unmodulated wave in place of the baseband signal in the reception mode.

The operation of the apparatus of this embodiment will be described below.
In the mobile terminal of the present embodiment, the transmission timing and the reception timing are shifted in time, and as shown in FIG. 9, the timing of the reception time slot is followed by the timing of the idle time slot, and then the transmission time slot. Then, the signal is transmitted / received by forming one cycle and repeating this cycle.

First, at the transmission timing, the transmission signal processing circuit 30 supplies the transmission baseband signal to the quadrature modulator 31. Therefore, the quadrature modulator 31 outputs π / 4.
A shift QPSK modulated wave is output, and this modulated wave is supplied to the transmission unit 32 through the switch 517. At this time, the quadrature modulator 31 uses the 9th signal from the PLL synthesizer circuit 50 ″.
Modulation is performed using a 50 MHz local oscillation signal.

Next, at the reception timing, the transmission signal processing circuit 30 supplies a DC voltage to the quadrature modulator 31,
As a result, the output signal from the quadrature modulator 31 becomes an unmodulated wave. This unmodulated wave is supplied to the first frequency converter 47 through the switch 517 as a local oscillation signal of 950 MHz, and frequency conversion of the received RF signal to the first intermediate frequency is performed.

As described above, in the apparatus of the embodiment, the local oscillation signal for transmission and reception is shared by one signal, and this is switched by the switcher at the time of transmission and at the time of reception. A hybrid circuit that bifurcates a local oscillation signal and an amplifier that adjusts the level of each of the bifurcated oscillation signals are not required, and it is possible to reduce the power consumption of the amplifier, reduce the circuit scale, and reduce the cost.

The embodiment of FIG. 8 relates to a so-called direct modulation type communication terminal, but the present invention is not limited to this, and the present invention is applied to an indirect modulation type communication terminal. It may be.

FIG. 10 shows another embodiment of the present invention. In this embodiment, the mobile device shown in FIG. 18 described in the section of the prior art is improved to reduce power consumption, reduce size and weight, and reduce price, and is a TDMA communication device. In the figure, the difference from FIG. 18 is that the local oscillator 71 that supplies the oscillation signal for modulation to the modulator 53, which is conventionally required, is removed, and instead the oscillation signal of the first PLL synthesizer circuit 66 is directly modulated. That is, the signal is supplied to 53 as an oscillation signal for modulation. The first PLL synthesizer circuit 66 changes the oscillation frequency of the oscillation signal during transmission and reception, and the oscillation frequency of the output signal in each circuit is adjusted as follows.

The first PLL synthesizer circuit 66 outputs an oscillation signal having an oscillation frequency of 163 MHz during transmission and an oscillation frequency near 245 MHz during reception. However, it fluctuates about ± 200 KHz depending on the channel setting data. The second PLL synthesizer circuit 67 outputs an oscillation signal having an oscillation frequency near 1120 MHz. However, depending on the channel setting data ± 8
It fluctuates about MHz. The output signal of the frequency converter 68 and thus the distributor 69 is 1283 MHz ± 8 M in the transmission mode.
In the reception mode, the oscillation frequency is 1365 ± 8 MHz.

The operation principle of the apparatus of this embodiment will be described below.
In the mobile device adopting the TDMA system, the transmission time zone and the reception time zone are different as shown in FIG. 9, so by changing the oscillation frequency of the first PLL synthesizer circuit 66 between the transmission time zone and the reception time zone, 1st PLL during transmission
The synthesizer circuit 66 is also used as a generation source of a modulation oscillation signal supplied to the modulator 53.

FIG. 11 is a diagram for explaining the relationship between the frequencies at each point in the apparatus of this embodiment. Assuming that the oscillation frequency of the local oscillator 71 is f _M and the oscillation frequency of the first PLL synthesizer circuit 66 is f _p1 in the conventional circuit, the transmission time zone and the reception time zone are different as described above, and therefore the first PLL synthesizer circuit F _P1 when transmitting oscillation frequency, f when receiving
_{If P1} + f _M , the local oscillator 71 and the first PLL synthesizer 66 can be shared. Therefore, in the device of this embodiment, the oscillation frequency of the first PLL synthesizer circuit 66 is set to the oscillation output of the local oscillator 71 = 82 MHz, f _P1 = 163 MHz at the time of transmission, and f _P1 + f at the time of reception.
_M = 245 MHz. As a result, the frequency of the local oscillation signal supplied to the frequency converter 54 during transmission is 12
At 83 MHz, the frequency of the transmission RF signal output from the frequency converter 54 becomes 1446 MHz, while the frequency of the local oscillation signal supplied to the frequency converter 62 at the time of reception becomes 1365 MHz, which is the same as the conventional case.

In the apparatus of this embodiment, the first PLL synthesizer circuit 66 needs to rapidly change and stabilize the oscillation frequency at the time of transmission and at the time of reception, and therefore has a circuit configuration different from the conventional circuit configuration. There is. Figure 12 shows this first P
It is one configuration example of the LL synthesizer circuit 66. The difference from the conventional configuration is that a capacitor Cp is connected in parallel to a variable capacitance diode VD in a voltage controlled oscillator circuit 664 via a switch SW, and this switch SW is closed during transmission and opened during reception. With such a configuration, the oscillation signal is output by opening the switch SW at the time of reception in the same manner as the conventional circuit, while the switch SW is closed at the time of transmission to connect the capacitor Cp in parallel with the capacitor of the variable capacitance diode VD. As a result, the capacitance is increased and the resonance frequency of the LC resonance circuit (that is, the oscillation frequency of the output signal) is drastically lowered.

FIG. 13 shows another configuration example of the first PLL synthesizer circuit 66. In this example, on the anode side of the variable capacitance diode VD of the voltage controlled oscillation circuit 664,
The feature is that the DC voltage source Vp is connected through the switch SW. The switch SW is opened at the time of transmission and closed at the time of reception. With this configuration, the switch SW is opened during transmission to operate in the same manner as the conventional voltage controlled oscillator circuit, while the switch SW is closed during transmission to apply the DC voltage Vp to the anode side of the variable capacitance diode VD. As a result, the bias voltage applied to the variable capacitance diode VD is reduced, the capacitance of the diode VD is reduced, and the resonance frequency of the LC resonance circuit (that is, the oscillation frequency of the output signal) is rapidly increased.

As the control signal for the switch SW of the first PLL synthesizer circuit 66, it is not necessary to newly provide a control signal generating circuit, and this control data is added to the channel setting data from the conventional control circuit 51. Just include it.

The apparatus of the embodiment of FIG. 10 described above is the first PLL.
Although the frequency of the oscillation signal of the synthesizer circuit 66 is changed during transmission and reception, the present invention is not limited to this, and the oscillation frequency on the side of the first PLL synthesizer circuit 66 is fixed and the second PLL is used according to the same principle. The local oscillation frequency supplied to the frequency converters 54 and 62 may be switched between reception and transmission by changing the oscillation frequency of the synthesizer circuit 66.
That is, for example, the oscillation frequency of the first PLL synthesizer circuit 66 is 163 MHz (± 200 kHz), and the oscillation frequency of the second PLL synthesizer circuit 67 is 1120 MHz during transmission.
(± 8 MHz), switch to 1202 MHz (± 8 MHz) when receiving.

[0084]

As described above, according to the present invention,
It is possible to achieve low power consumption, small size, light weight, and low price of the mobile communication device.

[Brief description of drawings]

FIG. 1 is a diagram illustrating the principle of the present invention.

FIG. 2 is a diagram showing a mobile communication device as an embodiment of the present invention.

FIG. 3 is a diagram showing a configuration example of a drain setting circuit in the device of the embodiment.

FIG. 4 is a diagram showing input / output characteristics of a high-efficiency power amplifier in the device of the embodiment.

FIG. 5 is a diagram showing a mobile communication device as another embodiment of the present invention.

FIG. 6 is a diagram showing a circuit configuration example of an up-conversion mixer in the apparatus of the embodiment.

FIG. 7 is a diagram showing a circuit configuration example of a gain variable element in the apparatus of the embodiment.

FIG. 8 is a diagram showing a mobile communication device as another embodiment of the present invention.

FIG. 9 is a diagram illustrating a relationship between transmission / reception time of the device according to the embodiment.

FIG. 10 is a diagram showing a mobile communication device as still another embodiment of the present invention.

FIG. 11 is a diagram for explaining a frequency arrangement relationship in the device of the embodiment.

FIG. 12 is a diagram showing a circuit configuration example of a first PLL synthesizer circuit in the device of the embodiment.

FIG. 13 is a diagram showing another circuit configuration example of the first PLL synthesizer circuit in the embodiment apparatus.

FIG. 14 is an overall configuration diagram of a conventional digital mobile communication terminal.

FIG. 15 is a diagram showing a circuit configuration example of a high-efficiency power amplifier in a conventional device.

FIG. 16 is a diagram showing a circuit configuration example of a variable attenuator in a conventional device.

FIG. 17 is a diagram showing a configuration example of a PLL synthesizer circuit of a conventional device.

FIG. 18 is a diagram showing another configuration example (indirect modulation method) of a conventional digital mobile communication terminal.

FIG. 19 is a diagram showing a circuit configuration example of a PLL synthesizer circuit of a conventional device.

[Explanation of symbols]

 10 Display Operation Unit 20 Control Circuit 21 Tone Generation Circuit 22 Encoding / Decoding Circuit 23 Time Division Multiplexing / Demultiplexing Circuit 30 Transmission Signal Processing Circuit 31 Quadrature Modulator 311 Intermediate Frequency Quadrature Modulator 312 Up-Conversion Mixer 32 Transmitter 321 Variable Attenuation 322 High efficiency power amplifier 323 Gate voltage setting circuit 324 Drain voltage setting circuit 3241 Power control signal decoder 3242 Reference voltage source 3423 Error amplifier 3444 Transistor 3445 Feedback circuit 33 Transmitter / receiver duplexer 34 Gain variable element 40 Received signal processing circuit 41 Quadrature demodulation 42 limiter amplifier 43 intermediate frequency amplifier 44 second intermediate frequency filter 45 second frequency converter 46 first intermediate frequency filter 47 first frequency converter 48 band filter 49 low noise RF amplifier 50 PLL synthesizer times 511 1st PLL synthesizer circuit 512 2nd PLL synthesizer circuit 513 Frequency converter 514 Hybrid circuit 515, 516 Amplifier 52 Modulation baseband circuit 53 Modulator 54, 62, 63, 68 Frequency converter 55 Power amplifier 61 Low noise amplifier 64 Intermediate frequency amplifier 65 Demodulator 66, 67 PLL synthesizer circuit 69 Distributor 70, 71 Local oscillator

 ─────────────────────────────────────────────────── ─── Continuation of the front page (72) Hideki Konsago, Inventor Hideki Honago 1015, Kamiodanaka, Nakahara-ku, Kawasaki, Kanagawa, Fujitsu Limited

Claims (14)

[Claims] 1. A transmission unit for switching and setting transmission power in multiple stages according to a transmission power control signal, and a drain voltage to be supplied to a power amplifier constituting the transmission unit stepwise according to the transmission power control signal. A mobile communication device comprising: a drain voltage setting circuit which is switched to and set. 2. The mobile communication device according to claim 1, wherein the drain voltage setting circuit switches and sets the drain voltage only in several steps on the side of the higher transmission power in the transmission power control. 3. The mobile communication device according to claim 1, further comprising a gate voltage setting circuit that sets a gate voltage to be supplied to a power amplifier that constitutes the transmitting unit in a stepwise manner according to the transmission power control signal. . 4. The mobile communication device according to claim 1, wherein when the power amplifier has a multi-stage amplifier configuration, the drain voltage setting circuit is provided only in an amplification stage having a large inflow current. 5. The mobile communication device according to claim 1, wherein when the power amplifier has a multi-stage amplifier configuration, each amplification stage is provided with a drain voltage setting circuit. 6. A modulator for modulating a transmission baseband signal to an intermediate frequency band by a predetermined modulation method, a frequency converter for converting a modulated wave output of the modulator to a desired radio transmission wave frequency, and the frequency conversion. Oscillation source for generating a local oscillation signal used for frequency conversion in a power supply, and a gain to be supplied to the frequency conversion section by switching the gain of the local oscillation signal of the local generation source stepwise according to the transmission power control signal A mobile communication device including a variable element. 7. A second gain variable element is provided before or after the frequency converter, and the gain of the gain variable element is switched stepwise according to the transmission power control signal, and the gain variable element and the second variable gain element are provided. 7. The mobile communication device according to claim 6, wherein the transmission power is switched in multiple stages by a combination of the variable gain elements. 8. The structure according to claim 6, wherein the amplitude value of the transmission baseband is switched stepwise according to the transmission power control signal, and the combination of this and the gain variable element switches the transmission power in multiple steps. The mobile communication device described. 9. A drain voltage setting circuit for setting the drain voltage supplied to a power amplifier for power amplification of the output signal of the frequency converter in a stepwise manner according to the transmission power control signal. The mobile communication device described. 10. The mobile communication device according to claim 6, wherein the variable gain element is controlled to a maximum attenuation state or an operation stop state when transmission is not performed. 11. A modulator for modulating a transmission baseband signal by a predetermined modulation method, a local oscillating circuit for generating a local oscillating signal used for a modulation operation in the modulator, and supplying the local oscillating signal to the modulator. Equipped with a switch that switches the output signal of the modulator to the transmission power amplifier side at the time of transmission and to the frequency converter side of the reception circuit at the time of reception, and stops the transmission of the transmission baseband signal to the modulator at the time of reception. And a mobile communication device configured so that the modulator output is an unmodulated wave. 12. A mobile communication device for performing communication by shifting a transmission timing and a reception timing, the first and second PLL synthesizer circuits, and frequency conversion from these output signals to radio transmission waves and radio reception waves. And a frequency conversion circuit for generating a combined local oscillation signal for performing intermediate frequency conversion of the first PLL synthesizer circuit, and an oscillation output signal of the first PLL synthesizer circuit as a modulation local oscillation signal of the transmission side modulator. And first,
The frequency of the oscillation output signal of any one of the second PLL synthesizer circuits is changed at the time of transmission and at the time of reception, so that the synthesis local unit is adapted to the frequency conversion of the radio transmission wave and the intermediate frequency conversion of the radio reception wave. A mobile communication device configured to change the frequency of an oscillation signal. 13. A PLL synthesizer circuit that changes the frequency during transmission and reception adds a capacitor in parallel with a variable capacitance diode that changes the oscillation frequency of a voltage controlled oscillator circuit, and switches the capacitor and the variable capacitance diode. The mobile communication device according to claim 12, wherein the means is configured to connect or disconnect at the time of transmission and at the time of reception. 14. A PLL synthesizer circuit that changes a frequency during transmission and reception includes an applying circuit that applies a bias voltage to a variable capacitance diode that changes an oscillating frequency of a voltage controlled oscillating circuit. 13. The mobile communication device according to claim 12, wherein the voltage is changed between transmission and reception.